This invention relates to a distortion compensation device, and in particular a distortion compensation device comprising a function which determines the delay time occurring in the feedback loop with the power amplifier from the SNR (signal-to-noise ratio), the ACLR (adjacent channel power ratio), the noise level Pn or similar, and based on the delay time controls the timing within each portion of the distortion compensation device.
In recent years bandwidth resources have become restricted, and there has been widespread adoption of highly efficient digital transmission in wireless communications. When employing multivalued phase modulation methods to wireless communication, techniques to linearize the amplification characteristics of the transmitting side and particularly the power amplifier to suppress nonlinear distortion, and to reduce adjacent channel leakage power, are important; further, when using amplifiers with poor linearity to improve power efficiency, techniques to compensate for the occurrence of distortion arising therefrom are essential.
FIG. 26 is a block diagram showing one example of a transmission device in conventional wireless equipment; the transmission signal generation device 1 transmits a serial digital data stream, and the serial/parallel converter (S/P converter) 2 converts the digital data stream into two series, which are an in-phase component (I signal) and quadrature component (Q signal), distributing alternating bits. The DA converter 3 converts the I signal and Q signal into analog baseband signals, for input to the quadrature modulator 4. The quadrature modulator multiplies the input I and Q signals (transmission baseband signals) by the reference carrier and a signal phase-shifted 90° from this, respectively, and adds the multiplied results to perform quadrature conversion for output. The frequency converter 5 mixes the quadrature-modulated signal and the local oscillation signal to perform frequency conversion, and the transmission power amplifier 6 amplifies and radiates through the air, from an aerial wire (antenna) 7, the carrier wave output from the frequency converter 5.
In W-CDMA, PDC (Personal Digital Cellular) and other mobile communications, the transmission power of the transmission device is high, ranging from 10 mW to several tens of watts, and the input/output characteristic (distortion function f(p)) of the transmission power amplifier 6 are nonlinear, as indicated by the dashed line in (a) in FIG. 27. Nonlinear distortion occurs due to this nonlinear characteristic, and the frequency spectrum in the vicinity of the transmission frequency f0 has lifted side lobes, as indicated by the solid line in (b) in FIG. 27, so that leakage into adjacent channels and adjacent channel interference occur. That is, due to nonlinear distortion, the power leaked by the transmission wave into adjacent frequency channels is large as shown in (b). The leakage power is explained as the ACLR (Adjacent Channel Leakage Ratio), but normally is the same as the ACPR (Adjacent Channel Power Ratio). The ACLR is the ratio of the power of the channel in question, which is the area of the spectrum between the dot-dash line A and the dot-dash line A′ in (b) of FIG. 27, to the adjacent leakage power which is the area of the leakage spectrum in the adjacent channels between the two dot-dash lines B, B′. This leakage power is noise with respect to other channels, and causes degradation of the communication quality of these channels. Hence it must be strictly regulated.
The leakage power is for example low in the linear region of the power amplifier (see (a) in FIG. 27), and increases in the nonlinear region. In order to enable use as a high-output transmission power amplifier, the linear region must be broadened. However, in order to achieve this an amplifier is required with capacity equal to or greater than that actually required, so that there are problems of disadvantageous cost and device size.
Further, in normal amplifiers the power added efficiency in the linear region is low, as seen in FIG. 28, and is large in the nonlinear region. Here the “power added efficiency” is the ratio of the difference in the input power Pin and output power Pout (Pin-Pout) to the rated power of the amplifier (expressed in percent), and is the portion which becomes heat. Thus in order to obtain the required transmission power, a large power consumption is necessary in the nonlinear region, and yet as described above, distortion increases and the ACLR is degraded. In such circumstances, a wireless device with a distortion compensation function (a linearizer) compensates for distortion in the transmission power and enables use of the amplifier in the region with good power added efficiency.
FIG. 29 is a block diagram of a transmission device comprising a digital nonlinear distortion compensation function using a DSP. The digital data group (transmission signal) transmitted from the transmission signal generation device 1 is converted into two series, an I signal and a Q signal, in the S/P converter 2, and these are input to the distortion compensation portion 8 comprising a DSP. As shown functionally in FIG. 30, the distortion compensation portion 8 comprises a distortion compensation coefficient storage portion 8a, which stores distortion compensation coefficients h(pi) (i=0 to 1023) according to the power levels 0 to 1023 of the transmission signal x(t), a predistortion portion 8b which performs distortion compensation processing (predistortion) of the transmission signal using the distortion compensation coefficients h(pi) according to the transmission signal level, and a distortion compensation coefficient calculation portion 8c which compares the transmission signal x(t) with a demodulation signal (feedback signal) y(t) demodulated by a quadrature detector, described below, calculates the distortion compensation coefficients h(pi) such that the difference is zero, and performs updating.
The distortion compensation portion 8 uses the distortion compensation coefficients h(pi) according to the power level of the transmission signal x(t) to perform predistortion processing of the transmission signal, and inputs the result to the DA converter 3. The DA converter 3 converts the input I signal and Q signal into analog baseband signals, and inputs the result to the quadrature modulator 4. The quadrature modulator 4 multiplies the I signal and Q signal thus input by the reference carrier wave and by the signal obtained by phase-shifting this reference carrier wave by 90°, respectively, and by adding the multiplication results performs quadrature conversion for output. The frequency converter 5 mixes the quadrature modulated signal and the local oscillation signal to perform frequency conversion, and the transmission power amplifier 6 power-amplifies the carrier wave signal output from the frequency converter 5, and radiates the result through the air, from an aerial wire (antenna) 7.
A portion of the transmission signal is input to the frequency converter 10 via a directional coupler 9, and is frequency-converted and input to the quadrature detector 11. The quadrature detector 11 multiplies the input signals by the reference carrier wave and by the signal resulting from 90° phase-shifting of the carrier wave respectively to perform quadrature detection, and the baseband I and Q signals are reproduced on the transmission side and input to the AD converter 12. The AD converter 12 converts the I and Q signals input into digital signals, which are input to the distortion compensation portion 8. The distortion compensation portion 8 uses the LMS (least-mean-square) algorithm in adaptive signal processing to compare the transmission signal prior to distortion compensation with the feedback signal resulting from demodulation by the quadrature detector 11, and calculates and updates the distortion compensation coefficients h(pi) such that the difference is zero. Then, the distortion compensation coefficients updated for the transmission signal to be transmitted next are used to perform predistortion processing and output. Subsequently the above-described operation is repeated, so that the nonlinear distortion of the transmission power amplifier 6 is suppressed and adjacent-channel leakage power is reduced.
FIG. 31 is an explanatory diagram of distortion compensation processing using an adaptive LMS algorithm. 15a is a multiplier (corresponding to the predistortion portion 8b in FIG. 30) which multiplies the transmission signal x(t) by the distortion compensation coefficients hn−1(p), 15b is a transmission power amplifier having a distortion function f(p), 15c is a feedback system which feeds back the output signal y(t) from the transmission power amplifier, 15d is a calculation portion (amplitude-power conversion portion) which calculates the power p (=x(t)2) of the transmission signal x(t), and 15e is a distortion compensation coefficient storage portion (corresponding to the distortion compensation coefficient storage portion 8a in FIG. 30) which stores distortion compensation coefficients for different power levels of the transmission signal x(t); distortion compensation coefficients hn−1(p) are output according to the power p of the transmission signal x(t), and in addition the distortion compensation coefficients hn−1(p) are updated with the distortion compensation coefficients hn(p) determined by the LMS algorithm.
15f is a complex conjugate signal output portion, 15g is a subtracter which outputs the difference e(t) between the transmission signal x(t) and the feedback demodulation signal y(t), 15h is a multiplier which performs multiplication of e(t) and u*(t), 15i is a multiplier which performs multiplication of hn−1(p) and y*(t), 15j is a multiplier which multiplies the step size parameter R, 15k is an adder which adds hn−1(p) and Re(t)u*(t), and 15m, 15n, and 15p are delay portions, which append to the input signal the delay time from the input of the transmission signal x(t) until the feedback demodulation signal y(t) is input to the subtracter 15g. 15f and 15h through 15j constitute a rotation calculation portion 16. u(t) is the signal receiving the distortion. As a result of the above configuration, the following calculations are performed.
 hn(p)=hn−1(p)+μe(t)u*(t)e(t)=x(t)−y(t)y(t)=hn−1(p)x(t)f(p)u(t)=x(t)f(p)=h*n−1(p)y(t)P=|x(t)|2
Here x, y, f, h, u, e are complex numbers, and * indicates the complex conjugate. By performing the above-described calculations, distortion compensation coefficients h(p) are updated such that the difference e(t) between the transmission signal x(t) and the feedback demodulation signal y(t) is minimum, so that ultimately there is convergence on the optimum distortion compensation coefficient values, and the distortion of the transmission power amplifier is compensated.
FIG. 32 shows the overall configuration of a transmission device represented as x(t)=I(t)+jQ(t); portions which are the same as in FIG. 29 and FIG. 31 are assigned the same symbols.
As described above, the principle of digital nonlinear distortion compensation methods involves feedback detection of the carrier wave obtained by quadrature modulation, digital conversion and comparison of the transmission signal and feedback signal, and realtime updating of the distortion compensation coefficients based on the comparison results. By means of this nonlinear distortion compensation method, distortion can be reduced, and consequently leakage power can be kept low even at high-output operation in the nonlinear range; in addition, the power added efficiency can be improved.
However, if the delay time in the transmission power amplifier 15b is D0 and the delay time in the feedback system 15c is D1, then the delay time D set in each of the delay circuits 15m, 15n, 15p of the distortion compensation device must be set so as to satisfy the equation:D=D0+D1
If this delay time D cannot be correctly set, the distortion compensation function will not operate effectively, and the larger the error of the delay time setting, the greater the swelling of the side lobes, and the more pronounced the power leakage into adjacent channels.
There are individual differences and changes with aging in the filters and other devices used in the power amplifier 15b and feedback system 15c, so that the overall delay time D fluctuates. Consequently the delay times set in the delay circuits 15m, 15n, 15p for each conventional transmission device are adjusted manually using delay time adjustment switches or similar, and so the task of adjusting delay times is complex, and there is the further problem that precise adjustments cannot be made.